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Old 27th Feb 2019, 1:06 pm   #21
Skywave
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Arrow Re: Q-meter wanted

Quote:
Originally Posted by Ed_Dinning View Post
The best and simplest Q meter for the OP's application is the Advance T2. It is simple to use and is fitted with calibrated variable C and goes from 50KHz to 100MHz.
The T2 that I have here states that the lowest freq. it will tune to is 100 kHz, although the scale is calibrated down to 70 kHz.
I am in agreement with all the other comments in that post from Ed.

Al.
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Old 27th Feb 2019, 1:27 pm   #22
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Default Re: Q-meter wanted

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Originally Posted by saddlestone-man View Post
The Electronics Australia, June 1969 Q-meter design looks very useful, and easily reproducible today. The full article covers six pages and a scan of this issue of the magazine can be found at: https://www.americanradiohistory.com...stralia_AU.htm
That is handy that you found a source for the article already fully scanned.

Actually, that particular month of Electronics Australia also had a lot of other interesting articles. One was about LCD flat panel TV developed by RCA but also, on page 98, the best ever Solid State Theremin designed by non other than Leo Simpson, who became the editor of Silicon Chip. But if that was not enough the brilliant spoof article on Calibrated Crickets and a really interesting article on speakers that provide electro-mechanical amplification. The issue was a gem.
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Old 27th Feb 2019, 1:58 pm   #23
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Default Re: Q-meter wanted

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Originally Posted by saddlestone-man View Post
The Electronics Australia, June 1969 Q-meter design looks very useful, and easily reproducible today. The full article covers six pages and a scan of this issue of the magazine can be found at: https://www.americanradiohistory.com...stralia_AU.htm
Thanks. I had a quick skim through it over lunch and it's probably worth mentioning that the source impedance for that Q meter is extremely high at either 4.7R or 0.9R. The author does offer equation 10 to correct for this but I think the impact of the high source impedance is a bit underplayed in the article.

Without using equation 10 after the measurement the meter design is seriously flawed if the constructor is expecting to make a 30MHz Q meter capable of measuring medium to high Q inductors. Otherwise, huge errors can occur with certain inductors.

The old school lab Q meters like the TF1245 and the HP4342A strive for errors within +/-10% and often the error will be lower than this. For certain inductors at certain frequencies I'd expect to see errors of maybe 50% with the Rowe Q meter unless equation 10 is used.

To give a comparison, one of my lashup Q jigs has a source impedance of 0.01 ohm. About two orders of magnitude lower and the jig cost pennies to make. I suppose the source impedance of the Rowe meter could be improved with the use of modern and accurate resistors.

I've not explored the other limitations of the Rowe meter yet but I just want to walk away from it. By all means build one because it is cheap to make but it has very limited design integrity in my opinion.
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Old 27th Feb 2019, 3:53 pm   #24
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Default Re: Q-meter wanted

Hi Jeremy

Can you publish here your jig which produces this very low drive impedance, presumably over a broad frequency range. As you say this is the weak point of simple Q-meters, including the 'Australian' one.

best regards ... Stef
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Old 27th Feb 2019, 4:48 pm   #25
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Default Re: Q-meter wanted

It's really just a couple of chip resistors but I should add that I don't use the E/e method with this jig because the 0.01R resistance of these cheapo SMD resistors won't be accurate at RF frequencies. In this case, the resistance just has to remain very low compared to the ESR of the inductor under test. I use the swept -3dB method to determine Q with this jig rather than the E/e method that most Q meters use. So the accuracy of the 0.01R source isn't critical. It just has to behave itself and remain a low Z source up to maybe 30MHz.

I have another jig that creates a low Z source using a special transmission line section and this works up into VHF and I also use the -3dB method here.

I also use the series resonance method in a controlled impedance jig and this measures Q with respect to the shunt loss of the series circuit. This method doesn't use a 0.01R resistance anywhere.

These jigs are all cheap to make and mine look scruffy and tired but they work well Both these methods are slow compared to a Q meter because they require some computation but I was going to see if it is possible to make something with an AD2 that reads Q directly on a PC screen.

I think most people can get by with the swept -3dB method or the insertion loss method. I think both of these methods could be done with an AD2 but I haven't tried it. I also use E and H field probes to help measure Q and I get good results with this but this isn't really suitable down at LW and MW.
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Old 27th Feb 2019, 5:42 pm   #26
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Default Re: Q-meter wanted

Gents
Thank you for the contributions. I have a quick scan at the Electronics Austrialia article June 1969. Equation 6 and 7 are identical to equation 3.9 and 3.10 in the Section 3-41 to 3-47 of the HP4342 Q-meter's manual:

http://www.docente.unicas.it/userupl...ilent_4342.pdf

The measurement of the stray capacitance of an inductance and the effect of oscillator frequency close to the self resonance frequency SRF got me intrigued for a while. I have been trying to dig deeper into the orgins of those equations but no one seemed to know; here they appear again.

Someone told me the variable cap of the HP4342 is gold plated. That adds to my Q-meter fever.

Last edited by regenfreak; 27th Feb 2019 at 5:49 pm.
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Old 27th Feb 2019, 9:59 pm   #27
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Originally Posted by regenfreak View Post
Someone in another forum had suggested me to download the manual and look at the method used in HP4342 instructions of measuring the stray capacitance of an inductor . But the method had turned out to not useful. But I must say the HP4342 manual has one or two interesting graphs and I was intrigued by them.

The measurement of the stray capacitance of an inductance and the effect of oscillator frequency close to the self resonance frequency SRF got me intrigued for a while. I have been trying to dig deeper into the orgins of those equations but no one seemed to know; here they appear again.
I downloaded the graphs from the HP manual and I think you may mean the graphs below.

I tried to conjure up a worked example that ties in with the graphs.

If you had a 'test network' that was modelled by 100uH in series with 17.4 ohms and there was 8pF across it (all at F = 2.77MHz) I think the HP4342A would tell you that this is a 132uH inductor with a Q of 75 when it tests it at 2.77MHz. The HP4342A would need to be set to 25pF to peak up the Q reading and it would peak with the Q at about 75.

I think the goal of all those procedures and equations is to take you through the process of reverse engineering from "132uH inductor with a Q of 75" to creating a better model of the network that would consist of a 100uH inductor with a Q of 100 (i.e. Rs =17.4R) but where someone has then added 8pF across it.

One part of the procedure is to measure the self resonance of the coil in the test network using the HP4342A procedures for that. i.e. fit a suitable reference inductor and find a test frequency where the addition of this test network only affects the Q of the reference inductor and not its reactance. In this case this would happen at around 5.6MHz. This what HP would refer to as the self resonant frequency Fo.

If the original test frequency was 2.77MHz then F/Fo = 2.77/5.6 = 0.5

So look at the circle in the top graph. This is at 0.5 and shows that the Q correction factor is 0.75. So this tells you the factor the Q has fallen by a factor of 0.75 because of the parallel 8pF. So then hop down to the graph below and look at where 0.75 and 25pF intersect and it is 8pF.

So this predicts the network (at 2.77MHz) that read 132uH with a Q of 75 actually consists of a 100uH inductor with a corrected Q of (75/0.75) = 100 but where someone has then added 8pF across it.

This corrected network would be a better model to use in a simulator for example. However, this is very dated (and flawed) method for accurately modelling inductors up near resonance.
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Old 27th Feb 2019, 10:47 pm   #28
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Sorry, I forgot to include the step that produces the derating factor for the inductor from the top graph. I think you are supposed to divide the 132uH by the value in the top graph associated with F/Fo = 0.5 in this case. So this would be just over 1.25 according to the green circled area in the top graph. So the corrected L would be 132/1.26 = 104.8uH. This is close enough to 100uH I think.

This all assumes I understand the goal of all those graphs! I've not tried putting these numbers in to the equations to see if it all agrees.

I suppose the other thing to consider is that the top graph only scales to an F/Fo of 0.5. This tells us that this crude modelling method won't be very accurate if you start venturing too close to Fo with the Q meter. Fo would be 5.6MHz in this case. This is because you can't really produce an accurate wideband model with a simple fixed/lumped capacitance (that represents the 'distributed capacitance of the inductor) if you then want to explore the model closer and closer to what HP refer to here as the resonance frequency Fo.
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Old 27th Feb 2019, 11:26 pm   #29
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Brilliant Jeremy many thanks for very clear example. Now I understand what they mean. I was staring at those two graphs for a long time last week...what really intrigued me was the indicated inductance graph. I guess max f/f0=0.5 is the design operation limit for the Q meter. I was wondering what the graph would look like when f/fo > 0.5 and why the Q-meter was not designed to test at closer to fs, so I used a known inductor of 100micrHenries with silver plated variable caps, driving the LC tank with a signal generator and then used the resonance frequency equation to find the ratio between indicated inductance and actual(corrected) inductance. Here is my measurement result in the attachment below. It looks very much like the top graph you have discussed in HP4342 manual.

The reason I have been obsessive about measuring the stray capacitance is that I have been digging deep into the superheterodyne tracking design methods for a few weeks. I am designing and building my first valve superheterodyne radio from scratch ( I consider myself an outsider in the world of radio; I only started electronics as a hobby 1.5 years ago and never owned a working valve radio or repaired a radio in my life). This will involve designing and winding the my oscillator and RF coils; and eventually winding my own IF tranformers. After much thinking, I have realised that the only way to produce minimum tracking error solutions is to measure the actual stray capacitance of both the local oscillator and RF tank coils, and then re-calculate the three-frequency tracking analytical solutions. In most radio textbooks, the stray capacitance values of both local oscillator and RF tank are usually assumed values in the tracking calculations which will lead to non-negligible tracking errors.

Going back to the HP4342A manual section 3.45, If you go through the steps from (a) to (k), leading to equation (3.4) and (3.7). I am interested in the derivation and assumptions of these two simple equations (3.4 and 3.7) because I can see something "funny" about those steps: I got very confused about step (e) to (h). At step (e), you increase the frequency 10 x f1 and then swap the supplement inductor, and add the test inductor again...


I asked about this in another forum but nobody seems to know why the stray capacitance is measured and calculated in that particular manner in the manual. You see Cs is calculated and measured differently when Cs>10pF in section 3.47.
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Old 28th Feb 2019, 12:36 am   #30
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I think I have found the partial answer of my questions...Tehman 1943 p992, figure 28,

assuming f is inversely proportional to square C

The other equation is to do with the fundamental frequency and second harmonics

http://www.tuks.nl/pdf/Reference_Mat...20-%201943.pdf
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Old 28th Feb 2019, 12:41 am   #31
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I'll try and find some time to look deeper but if I had a HP4342A and wanted to use that procedure here I'd use something like this hasty excel sheet image below. This uses HP's equations from the HP4342A manual.

Note that their equation 3-7 is a bit dodgy on their page 3-15. For C1 I think they mean to refer to an earlier 'step b' that is the (combined) dial capacitance when initially testing the inductor. i.e. C1 should be the dial capacitance C at step b in 3-40. They also have two equations labelled 3-7. I must admit I cheated a bit because I skimmed a lot of it and just plucked the bits I was expecting to see and the result is the excel sheet below.

At least some of their equations do seem to work because I get good agreement with my own analysis of the same example network and my RF simulator also agrees very well.
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Old 28th Feb 2019, 12:59 am   #32
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Default Re: Q-meter wanted

Quote:
Originally Posted by regenfreak View Post
I have been digging deep into the superheterodyne tracking design methods for a few weeks. I am designing and building my first valve superheterodyne radio from scratch (
I have also always sought for minimum tracking errors in my home made superhets. The issues crop up using a variable capacitor for the RF (or antenna) tuning with the same value as the osc section. It becomes a matter of calculating the padder capacitor accurately. The method I have used includes that outlined in Terman's book. Later I found there were software utilities on the net that were quicker (reduced the chances of an error in the arithmetic).

Of course as you will be aware, the tracking can only be bang on at three points in the tuning range, in the middle and near the ends. So to average the errors its best that the zero tracking error points on either side of the band are inside the tuned range, not on the edge of it. Also the wider the tuned range the greater the absolute errors. In my shortwave radios they tune from 5.7 to 19 MHz in one sweep so it important that the errors are minimized with exactly the right padder value. If that range is broken into 3 separate bands for example, its much easier.

The whole problem was largely improved with special variable capacitors which have a smaller section for the oscillator gang and don't require a padder, like most (but not all) transistor radios have.

Once the calculations are done though, it still pays to check that the worst case tracking errors are equal and opposite in the tuned range on either side of center.
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Old 28th Feb 2019, 1:16 am   #33
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Quote:
I have also always sought for minimum tracking errors in my home made superhets. The issues crop up using a variable capacitor for the RF (or antenna) tuning with the same value as the osc section. It becomes a matter of calculating the padder capacitor accurately. The method I have used includes that outlined in Terman's book. Later I found there were software utilities on the net that were quicker (reduced the chances of an error in the arithmetic).
Which online calculator you are using? I have spotted a major bug in one of them; I have been talking to the guy who developed the calculator. He didn't know until I pointed it out:

http://electronbunker.ca/eb/Downloads.html

The solutions from the version available form his web site are not correct. Although he said he is trying to fix the bug but it has turned to be very complex problem.

I have used Langford-Smith method and also been looking at Henney's method.

My problem is easier as I am building a MW superhet but I have found the limitations of the two online calculators that I have been poking around...none of them can include the stray capacitance or calculate them correctly in the software.

This tracking calculator is not accurate because it ignores the stray capacitance completely:
http://tubularelectronics.com/?p=70

Last edited by regenfreak; 28th Feb 2019 at 1:27 am.
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Old 28th Feb 2019, 1:35 am   #34
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Thanks Jeremy



Now I know where those equations coming from.

It looks like I can measure stray capacitance without a Q-meter;

Simply plotting 1/f2 versus C;

the y-intercept is 1/fo2, negative x-intercept is Co, the slope = 4 x (pi)2xL
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Old 28th Feb 2019, 1:36 am   #35
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Which online calculator you are using? I have spotted a major bug in one of them; I have been talking to the guy who developed the calculator. He didn't know until I pointed it out.

I have used Langford-Smith method and also been looking at Henny's method.

My problem is easier as I am building a MW superhet but I have found the limitations of the two online calculators that I have been poking around...none of them can include the stray capacitance or calculate them correctly in the software
I can't point to the exact one I used, it was 4 years ago when I built these two MW/SW radios with a wide range SW band:

http://worldphaco.com/uploads/THE_EF98-OC16_RADIO.pdf

http://worldphaco.com/uploads/WORLDFETRON.pdf

Though, I checked the calculations manually and they closely agreed with Terman's method.

The thing is, regardless of calculations getting you 90% or more there, the exact value may need to be adjusted to allow for stray capacitances. So you really have to measure the tracking errors when it is done and plot that in a way to see if the errors are as evenly averaged over the entire band as possible.

As I recall I did that by having a frequency counter with an IF offset on the L/O (my radios all have buffered L/O outputs) and then injecting that f value into the RF circuits and deactivating the L/O and finding if the RF tuning was peaked or not, with a detector circuit added, & measuring the offset if it was not, by re-activating the L/O with the V/C in its new position and getting frequency error data that way representing the difference between the received frequency and the frequency that the RF stages were actually tuned to.Though there may be better methods.
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Old 28th Feb 2019, 1:53 am   #36
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Originally Posted by G0HZU_JMR View Post
Note that their equation 3-7 is a bit dodgy on their page 3-15. For C1 I think they mean to refer to an earlier 'step b' that is the (combined) dial capacitance when initially testing the inductor. i.e. C1 should be the dial capacitance C at step b in 3-40. They also have two equations labelled 3-7. I must admit I cheated a bit because I skimmed a lot of it and just plucked the bits I was expecting to see and the result is the excel sheet below.

At least some of their equations do seem to work because I get good agreement with my own analysis of the same example network and my RF simulator also agrees very well.
Equation 3-7 is derived from 1/f2 is proportional to C

The simplified equation 3.7 is at low frequency when f<<fo. The -1.0 is comparatively small and is therefore ignored in the denominator.

Now there is a design limit for f in a Q meter. It has become a problem if f > 0.5fo. I think those supplemental inductors are "fixes" to patch the design limit?

Last edited by regenfreak; 28th Feb 2019 at 2:14 am.
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Old 28th Feb 2019, 2:01 am   #37
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I can't point to the exact one I used, it was 4 years ago when I built these two MW/SW radios with a wide range SW band:

http://worldphaco.com/uploads/THE_EF98-OC16_RADIO.pdf

http://worldphaco.com/uploads/WORLDFETRON.pdf

Though, I checked the calculations manually and they closely agreed with Terman's method.

The thing is, regardless of calculations getting you 90% or more there, the exact value may need to be adjusted to allow for stray capacitances. So you really have to measure the tracking errors when it is done and plot that in a way to see if the errors are as evenly averaged over the entire band as possible.

As I recall I did that by having a frequency counter with an IF offset on the L/O (my radios all have buffered L/O outputs) and then injecting that f value into the RF circuits and deactivating the L/O and finding if the RF tuning was peaked or not, with a detector circuit added, & measuring the offset if it was not, by re-activating the L/O with the V/C in its new position and getting frequency error data that way representing the difference between the received frequency and the frequency that the RF stages were actually tuned to.Though there may be better methods.
Woww awesome they are beautifully built. Thanks for the ideas on how to measure the actual tracking errors..I am sure I will spend a lot of time figuring it out.

I just happen to be fascinated by the tracking problem...I actually find it very interesting otherwise I would cheat and not bother with it.
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Old 28th Feb 2019, 2:27 am   #38
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Now there is a design limit for f in a Q meter. It has become a problem if f > 0.5fo. I think those supplemental inductors are "fixes" to patch the design limit?
I'm not sure I'd describe the Q meter as having a design limit here. I think it is just the case that the corrected model it produces for C dist will have a creeping error with increasing frequency and the 'creep away' becomes quite noticeable above F/Fo of 0.5.

I think this is is an issue with the limitations of the model not the Q meter. If I used a VNA to try and predict a single fixed value for Cdist that works well over a wide bandwidth I think I'd end up with similar creep errors.

The lumped model for a solenoid inductor with Cdist is generally OK up to frequencies where the wire length used for the solenoid is a tiny fraction of a wavelength. A lot depends on the dimensions of the inductor but I think the model error for Cdist will begin to creep up when the inductor has a wire length of an eighth of a wavelength or so.

At work we abandoned this lumped type of model over 20 years ago unless it was used at frequencies well below these limits. But we mainly do ultra wideband design work so this model type falls apart big time for the stuff we do.
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Old 28th Feb 2019, 2:44 am   #39
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A while back I tested a 400nH solenoid for someone and modelled it with the works model. Back then I posted up a (silent) youtube video showing how well the model tracks reality. It's still on youtube and see the link below.

The tests are done up to frequencies about 4 times higher than the Fo frequency and the model does very well I think and it captures all of the resonance modes fairly accurately. This model is of an inductor with two terminals so it could be used as a component in a simulator for example.

Sorry the video is silent. I didn't have a microphone and I don't think my old version of camstudio works with audio on this PC anyway.

But the model tested below is a real model tested against a real inductor and not just an equation that predicts a theoretical capacitance to predict one spot resonance. The model predicts all kinds of resonances depending on the termination impedance at each end of the solenoid.

https://www.youtube.com/watch?v=4HSWW672vtc

To clarify things, the video is the equivalent of taking a real inductor and the model and being able to put each one in series on a simulated 2 port VNA between port 1 and 2. The simulated VNA can dynamically change its port 1 and port 2 impedances with the mouse. All the mouse does is change the VNA port impedances and the graphs all represent the processed data for things like insertion loss, complex impedance, group delay etc etc. The model and the real inductor agree very well I think. It isn't perfect above about 700MHz but still a very useful model and this is the basic version of the model.
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Old 28th Feb 2019, 3:33 am   #40
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thanks jeremy for sharing your expertise.
I didnt expect so much interest has been generated on the subject of Q meter and measurement.
Just a correction; the indicated inductance graph i plotted was obtained from a common FET oscillator that automatically oscillates at the resonace frequency of any parellel LC circuit. It was not driven by a signal generator.
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